Antenna structures and methods thereof for selecting antenna configurations

ABSTRACT

A system incorporating the subject disclosure may include, for example, a method including determining, by a communication device comprising a processor, a usage state of the communication device. The communication device includes selectable antennas, and the usage state includes an orientation of the communication device. The method also includes selecting a set of antennas according to the usage state, and obtaining an antenna gain pattern for an antenna in the selected set of antennas. The method further includes evaluating an expected performance of an antenna configuration corresponding to the selected set of antennas, relative to a performance of an existing antenna configuration. The method also includes initiating usage of the antenna configuration in accordance with improved performance relative to the existing antenna configuration. The antenna configuration comprises a polarization configuration, and the selected set of antennas comprises elements in different planes. Other embodiments are disclosed.

CROSS REFERENCE TO RELATED APPLICATIONS

The present application claims the benefit of priority to U.S. Provisional Application No. 61/896,233 filed on Oct. 28, 2013, which is hereby incorporated herein by reference in its entirety.

The present application claims the benefit of priority to U.S. Provisional Application No. 61/932,831 filed on Jan. 29, 2014, which is hereby incorporated herein by reference in its entirety.

The present application claims the benefit of priority to U.S. Provisional Application No. 61/941,888 filed on Feb. 19, 2014, which is hereby incorporated herein by reference in its entirety.

FIELD OF THE DISCLOSURE

The present disclosure relates generally to antenna structures and methods thereof for selecting antenna configurations, particularly configurations with multi-element antennas.

BACKGROUND

It is common for communications devices to have multiple antennas that are packaged close together (e.g., less than a quarter of a wavelength apart) and that can operate simultaneously within the same frequency band. Common examples of such communications devices include portable communications products such as cellular handsets, personal digital assistants (PDAs), and wireless networking devices or data cards for personal computers (PCs). Many system architectures (such as Multiple Input Multiple Output (MIMO)) and standard protocols for mobile wireless communications devices (such as 802.11n for wireless LAN, and 3G and 4G data communications such as 802.16e (WiMAX), HSDPA, 1xEVDO, and LTE) may require multiple antennas operating simultaneously.

BRIEF DESCRIPTION OF THE DRAWINGS

Reference will now be made to the accompanying drawings, which are not necessarily drawn to scale, and wherein:

FIG. 1A illustrates an antenna structure with two parallel dipoles;

FIG. 1B illustrates current flow resulting from excitation of one dipole in the antenna structure of FIG. 1A;

FIG. 1C illustrates a model corresponding to the antenna structure of FIG. 1A;

FIG. 1D is a graph illustrating scattering parameters for the FIG. 1C antenna structure;

FIG. 1E is a graph illustrating the current ratios for the FIG. 1C antenna structure;

FIG. 1F is a graph illustrating gain patterns for the FIG. 1C antenna structure;

FIG. 1G is a graph illustrating envelope correlation for the FIG. 1C antenna structure;

FIG. 2A illustrates an antenna structure with two parallel dipoles connected by connecting elements in accordance with one or more embodiments of the disclosure;

FIG. 2B illustrates a model corresponding to the antenna structure of FIG. 2A;

FIG. 2C is a graph illustrating scattering parameters for the FIG. 2B antenna structure;

FIG. 2D is a graph illustrating scattering parameters for the FIG. 2B antenna structure with lumped element impedance matching at both ports;

FIG. 2E is a graph illustrating the current ratios for the FIG. 2B antenna structure;

FIG. 2F is a graph illustrating gain patterns for the FIG. 2B antenna structure;

FIG. 2G is a graph illustrating envelope correlation for the FIG. 2B antenna structure;

FIG. 3A illustrates an antenna structure with two parallel dipoles connected by meandered connecting elements in accordance with one or more embodiments of the disclosure;

FIG. 3B is a graph showing scattering parameters for the FIG. 3A antenna structure;

FIG. 3C is a graph illustrating current ratios for the FIG. 3A antenna structure;

FIG. 3D is a graph illustrating gain patterns for the FIG. 3A antenna structure;

FIG. 3E is a graph illustrating envelope correlation for the FIG. 3A antenna structure;

FIG. 4 illustrates an antenna structure with a ground or counterpoise in accordance with one or more embodiments of the disclosure;

FIG. 5 illustrates a balanced antenna structure in accordance with one or more embodiments of the disclosure;

FIG. 6A illustrates an antenna structure in accordance with one or more embodiments of the disclosure;

FIG. 6B is a graph showing scattering parameters for the FIG. 6A antenna structure for a particular dipole width dimension;

FIG. 6C is a graph showing scattering parameters for the FIG. 6A antenna structure for another dipole width dimension;

FIG. 7 illustrates an antenna structure fabricated on a printed circuit board in accordance with one or more embodiments of the disclosure;

FIG. 8A illustrates an antenna structure having dual resonance in accordance with one or more embodiments of the disclosure;

FIG. 8B is a graph illustrating scattering parameters for the FIG. 8A antenna structure;

FIG. 9 illustrates a tunable antenna structure in accordance with one or more embodiments of the disclosure;

FIGS. 10A and 10B illustrate antenna structures having connecting elements positioned at different locations along the length of the antenna elements in accordance with one or more embodiments of the disclosure;

FIGS. 10C and 10D are graphs illustrating scattering parameters for the FIGS. 10A and 10B antenna structures, respectively;

FIG. 11 illustrates an antenna structure including connecting elements having switches in accordance with one or more embodiments of the disclosure;

FIG. 12 illustrates an antenna structure having a connecting element with a filter coupled thereto in accordance with one or more embodiments of the disclosure;

FIG. 13 illustrates an antenna structure having two connecting elements with filters coupled thereto in accordance with one or more embodiments of the disclosure;

FIG. 14 illustrates an antenna structure having a tunable connecting element in accordance with one or more embodiments of the disclosure;

FIG. 15 illustrates an antenna structure mounted on a PCB assembly in accordance with one or more embodiments of the disclosure;

FIG. 16 illustrates another antenna structure mounted on a PCB assembly in accordance with one or more embodiments of the disclosure;

FIG. 17 illustrates an alternate antenna structure that can be mounted on a PCB assembly in accordance with one or more embodiments of the disclosure;

FIG. 18A illustrates a three mode antenna structure in accordance with one or more embodiments of the disclosure;

FIG. 18B is a graph illustrating the gain patterns for the FIG. 18A antenna structure;

FIG. 19 illustrates an antenna and power amplifier combiner application for an antenna structure in accordance with one or more embodiments of the disclosure;

FIGS. 20A and 20B illustrate a multimode antenna structure useable, e.g., in a WiMAX USB or ExpressCard/34 device in accordance with one or more further embodiments of the subject disclosure;

FIG. 20C illustrates a test assembly used to measure the performance of the antenna of FIGS. 20A and 20B;

FIGS. 20D to 20J illustrate test measurement results for the antenna of FIGS. 20A and 20B;

FIGS. 21A and 21B illustrate a multimode antenna structure useable, e.g., in a WiMAX USB dongle in accordance with one or more alternate embodiments of the subject disclosure;

FIGS. 22A and 22B illustrate a multimode antenna structure useable, e.g., in a WiMAX USB dongle in accordance with one or more alternate embodiments of the subject disclosure;

FIG. 23A illustrates a test assembly used to measure the performance of the antenna of FIGS. 21A and 21B;

FIGS. 23B to 23K illustrate test measurement results for the antenna of FIGS. 21A and 21B;

FIG. 24 is a schematic block diagram of an antenna structure with a beam steering mechanism in accordance with one or more embodiments of the subject disclosure;

FIGS. 25A, 25B, 25C, 25D, 25E, 25F, and 25G illustrate a comparison of antenna pattern test measurement results;

FIG. 26 illustrates the gain advantage of an antenna structure in accordance with one or more embodiments of the subject disclosure as a function of the phase angle difference between feedpoints;

FIG. 27A is a schematic diagram illustrating a simple dual-band branch line monopole antenna structure;

FIG. 27B illustrates current distribution in the FIG. 27A antenna structure;

FIG. 27C is a schematic diagram illustrating a spurline band stop filter;

FIGS. 27D and 27E are test results illustrating frequency rejection in the FIG. 27A antenna structure;

FIG. 28 is a schematic diagram illustrating an antenna structure with a band-rejection slot in accordance with one or more embodiments of the subject disclosure;

FIG. 29A illustrates an alternate antenna structure with a band-rejection slot in accordance with one or more embodiments of the subject disclosure;

FIGS. 29B and 29C illustrate test measurement results for the FIG. 29A antenna structure;

FIG. 30 depicts an illustrative embodiment of an antenna structure in accordance with one or more embodiments;

FIG. 31 depicts an illustrative embodiment of a multiband antenna structure in accordance with one or more embodiments;

FIGS. 32A and 32B illustrate tuning using discrete selection of inductance to select antenna fundamental resonance frequency in accordance with one or more embodiments;

FIGS. 33A and 33B illustrate tuning using discrete selection of inductance to select fundamental resonance frequency where a separate but co-located high band element is shown with feed points F1H and F2H that allows for compatibility with RF transceiver front end designs requiring separate low- and mid- or low- and high-band connections to the antenna in accordance with one or more embodiments;

FIGS. 34A and 34B illustrate tuning and filtering using discrete selection of inductance to select antenna fundamental resonance frequency in accordance with one or more embodiments;

FIGS. 35A and 35B illustrate tuning and filtering using discrete selection of inductance to select fundamental resonance frequency in accordance with one or more embodiments;

FIG. 36 depicts an illustrative embodiment of polarization selectivity in a multi-antenna device having two antennas;

FIG. 37 depicts an illustrative embodiment of polarization selectivity in a multi-antenna device having two antennas in different planes;

FIG. 38 depicts an illustrative embodiment of polarization selectivity in a multi-antenna device including a multi-element antenna;

FIG. 39 depicts an illustrative embodiment of polarization selectivity in a multi-antenna device having four antennas;

FIG. 40 depicts an illustrative embodiment of polarization selectivity in a multi-antenna device including two multi-element antennas;

FIG. 41 depicts an illustrative embodiment of polarization selectivity in a multi-antenna device having eight antennas;

FIG. 42 depicts an illustrative embodiment of polarization selectivity in a multi-antenna device having a symmetric arrangement of four multi-element antennas;

FIG. 43 depicts an illustrative embodiment of polarization selectivity in a multi-antenna device having an asymmetric arrangement of four multi-element antennas;

FIG. 44 depicts an illustrative embodiment of polarization selectivity in a multi-antenna device having two movable antennas;

FIG. 45 depicts an illustrative embodiment of a method that can be applied to the subject disclosure;

FIG. 46 depicts an illustrative embodiment of a communication device; and

FIG. 47 is a diagrammatic representation of a machine in the form of a computer system within which a set of instructions, when executed, may cause the machine to perform any one or more of the methods described herein.

DETAILED DESCRIPTION

The subject disclosure describes, among other things, illustrative embodiments for a set of antennas for a communication device forming an antenna configuration having improved performance, where performance is evaluated by determining the envelope correlation coefficient (ECC) of the antenna configuration and antenna isolation. Other embodiments are described in the subject disclosure.

One embodiment of the subject disclosure includes a method comprising determining, by a communication device comprising a processor, a usage state of the communication device, wherein the communication device comprises a plurality of selectable antennas for engaging in wireless communications; the usage state includes an orientation of the communication device. The method also includes selecting a set of antennas from the plurality of antennas in accordance with the usage state, and obtaining an antenna gain pattern for an antenna in the selected set of antennas. The method further includes evaluating an expected performance of an antenna configuration corresponding to the selected set of antennas, relative to a performance of an existing antenna configuration. The method also includes initiating usage of the antenna configuration corresponding to the selected set of antennas in accordance with improved performance relative to the existing antenna configuration. The antenna configuration comprises a polarization configuration, and the selected set of antennas comprises a first antenna element in a first plane and a second antenna element in a different second plane.

One embodiment of the subject disclosure includes an antenna structure for a communication device comprising a set of antennas selected from a plurality of antennas of the communication device. The set of antennas is selected in accordance with a usage state of the communication device. An evaluation of performance of an antenna configuration corresponding to the selected set of antennas indicates improved performance relative to a previous antenna configuration; the evaluation includes an envelope correlation coefficient for the antenna configuration corresponding to the selected set of antennas and an antenna isolation for the antenna configuration corresponding to the selected set of antennas.

One embodiment of the subject disclosure includes a machine-readable storage medium comprising instructions. A processor, responsive to executing the instructions, performs operations. The operations comprise detecting a usage state of a communication device; the communication device comprises a plurality of selectable antennas for engaging in wireless communications, and the usage state includes an orientation of the communication device. The operations also comprise selecting a set of antennas from the plurality of antennas in accordance with the usage state, obtaining a first antenna gain pattern for a first antenna in the selected set of antennas, and obtaining a second antenna gain pattern for a second antenna in the selected set of antennas. The operations further comprise evaluating an expected performance of an antenna configuration corresponding to the selected set of antennas relative to a performance of an existing antenna configuration. The evaluating comprises determining an envelope correlation coefficient for the antenna configuration corresponding to the selected set of antennas and an antenna isolation for the antenna configuration corresponding to the selected set of antennas. The evaluating is based on the first antenna gain pattern and the second antenna gain pattern, and the second antenna gain pattern fits a null in the first antenna gain pattern.

Antenna structures in accordance with various embodiments of the disclosure are particularly useful in communications devices that require multiple antennas to be packaged close together (e.g., less than a quarter of a wavelength apart), including in devices where more than one antenna is used simultaneously and within the same frequency band or multiple frequency bands in cases where carrier aggregation is required. Common examples of such devices in which the antenna structures can be used include portable communications products such as cellular handsets, PDAs, smart phones, tablets, and wireless networking devices or data cards for PCs or other equipment integrated communication devices such as automobiles, trucks, or other vehicle categories. The antenna structures are also useful with system architectures such as MIMO and standard protocols for mobile wireless communications devices (such as 802.11n for wireless LAN, and 3G and 4G data communications such as 802.16e (WiMAX), HSDPA, 1xEVDO, LTE) that require multiple antennas operating simultaneously. The embodiments of the subject disclosure can be applied to future generations of wireless communication protocols such as 5G.

FIGS. 1A-1G illustrate the operation of an antenna structure 100. FIG. 1A schematically illustrates the antenna structure 100 having two parallel antennas, in particular parallel dipoles 102, 104, of length L. The dipoles 102, 104 are separated by a distance d, and are not connected by any connecting element. The dipoles 102, 104 have a fundamental resonant frequency that corresponds approximately to L=λ/2. Each dipole is connected to an independent transmit/receive system, which can operate at the same frequency. This system connection can have the same characteristic impedance z₀ for both antennas, which in this example is 50 ohms.

When one dipole is transmitting a signal, some of the signal being transmitted by the dipole will be coupled directly into the neighboring dipole. The maximum amount of coupling generally occurs near the half-wave resonant frequency of the individual dipole and generally increases as the separation distance d is made smaller. For example, for d<λ/3, the magnitude of coupling is greater than 0.1 or −10 dB, and for d<λ/8, the magnitude of the coupling is greater than −5 dB.

It is desirable to have no coupling (i.e., complete isolation) or to reduce the coupling (i.e., at least reduced isolation) between the antennas. If the coupling is, e.g., −10 dB, 10 percent of the transmit power is lost due to that amount of power being directly coupled into the neighboring antenna. There may also be detrimental system effects such as saturation or desensitization of a receiver connected to the neighboring antenna or degradation of the performance of a transmitter connected to the neighboring antenna. Currents induced on the neighboring antenna distort the gain pattern compared to that generated by an individual dipole. This effect is known to reduce the correlation between the gain patterns produced by the dipoles. Thus, while coupling may provide some pattern diversity, it has detrimental system impacts as described above.

Because of the close coupling, the antennas do not act independently and can be considered an antenna system having two pairs of terminals or ports that correspond to two different gain patterns. Use of either port involves substantially the entire structure including both dipoles. The parasitic excitation of the neighboring dipole enables diversity to be achieved at close dipole spacing, but currents excited on the dipole pass through the source impedance, and therefore manifest mutual coupling between ports.

FIG. 1C illustrates a model dipole pair corresponding to the antenna structure 100 shown in FIG. 1 used for simulations. In this example, the dipoles 102, 104 have a square cross section of 1 mm×1 mm and length (L) of 56 mm. These dimensions yield a center resonant frequency of 2.45 GHz when attached to a 50-ohm source. The free-space wavelength at this frequency is 122 mm. A plot of the scattering parameters S11 and S21 for a separation distance (d) of 10 mm, or approximately λ/12, is shown in FIG. 1D. Due to symmetry and reciprocity, S22=S11 and S12=S21. For simplicity, only S11 and S21 are shown and discussed. In this configuration, the coupling between dipoles as represented by S21 reaches a maximum of −3.7 dB.

FIG. 1E shows the ratio (identified as “Magnitude I2/I1” in the figure) of the vertical current on dipole 104 of the antenna structure to that on dipole 102 under the condition in which port 106 is excited and port 108 is passively terminated. The frequency at which the ratio of currents (dipole 104/dipole 102) is a maximum corresponds to the frequency of 180 degree phase differential between the dipole currents and is just slightly higher in frequency than the point of maximum coupling shown in FIG. 1D.

FIG. 1F shows azimuthal gain patterns for several frequencies with excitation of port 106. The patterns are not uniformly omni-directional and change with frequency due to the changing magnitude and phase of the coupling. Due to symmetry, the patterns resulting from excitation of port 108 would be the minor image of those for port 106. Therefore, the more asymmetrical the pattern is from left to right, the more diverse the patterns are in terms of gain magnitude.

Calculation of the correlation coefficient between patterns provides a quantitative characterization of the pattern diversity. FIG. 1G shows the calculated correlation between port 106 and port 108 antenna patterns. The correlation is much lower than is predicted by Clark's model for ideal dipoles. This is due to the differences in the patterns introduced by the mutual coupling.

FIGS. 2A-2F illustrate the operation of an exemplary two port antenna structure 200 in accordance with one or more embodiments of the disclosure. The two port antenna structure 200 includes two closely-spaced resonant antenna elements 202, 204 and provides both low pattern correlation and low coupling between ports 206, 208. FIG. 2A schematically illustrates the two port antenna structure 200. This structure is similar to the antenna structure 100 comprising the pair of dipoles shown in FIG. 1B, but additionally includes horizontal conductive connecting elements 210, 212 between the dipoles on either side of the ports 206, 208. The two ports 206, 208 are located in the same locations as with the FIG. 1 antenna structure. When one port is excited, the combined structure exhibits a resonance similar to that of the unattached pair of dipoles, but with a significant reduction in coupling and an increase in pattern diversity.

An exemplary model of the antenna structure 200 with a 10 mm dipole separation is shown in FIG. 2B. This structure has generally the same geometry as the antenna structure 100 shown in FIG. 1C, but with the addition of the two horizontal connecting elements 210, 212 electrically connecting the antenna elements slightly above and below the ports. This structure shows a strong resonance at the same frequency as unattached dipoles, but with very different scattering parameters as shown in FIG. 2C. There is a deep drop-out in coupling, below −20 dB, and a shift in the input impedance as indicated by S11. In this example, the best impedance match (S11 minimum) does not coincide with the lowest coupling (S21 minimum). A matching network can be used to improve the input impedance match and still achieve very low coupling as shown in FIG. 2D. In this example, a lumped element matching network comprising a series inductor followed by a shunt capacitor was added between each port and the structure.

FIG. 2E shows the ratio (indicated as “Magnitude I2/I1” in the figure) of the current on dipole element 204 to that on dipole element 202 resulting from excitation of port 206. This plot shows that below the resonant frequency, the currents are actually greater on dipole element 204. Near resonance, the currents on dipole element 204 begin to decrease relative to those on dipole element 202 with increasing frequency. A point of low coupling (2.44 GHz in this case) occurs near the frequency where currents on both dipole elements are generally equal in magnitude. At this frequency, the phase of the currents on dipole element 204 lag those of dipole element 202 by approximately 160 degrees.

Unlike the FIG. 1C dipoles without connecting elements, the currents on antenna element 204 of the FIG. 2B combined antenna structure 200 are not forced to pass through the terminal impedance of port 208. Instead a resonant mode is produced where the current flows down antenna element 204, across the connecting element 210, 212, and up antenna element 202 as indicated by the arrows shown on FIG. 2A. (Note that this current flow is representative of one half of the resonant cycle; during the other half, the current directions are reversed). The resonant mode of the combined structure features the following: (1) the currents on antenna element 204 largely bypass port 208, thereby allowing for high isolation between the ports 206, 208, and (2) the magnitude of the currents on both antenna elements 202,204 are approximately equal, which allows for dissimilar and uncorrelated gain patterns as described in further detail below.

Because the magnitude of currents is nearly equal on the antenna elements, a much more directional pattern is produced (as shown on FIG. 2F) than in the case of the FIG. 1C antenna structure 100 with unattached dipoles. When the currents are equal, the condition for nulling the pattern in the x (or phi=0) direction is for the phase of currents on dipole 204 to lag those of dipole 202 by the quantity π-kd (where k=2π/λ, and λ is the effective wavelength). Under this condition, fields propagating in the phi=0 direction from dipole 204 will be 180 degrees out of phase with those of dipole 202, and the combination of the two will therefore have a null in the phi=0 direction.

In the model example of FIG. 2B, d is 10 mm or an effective electrical length of λ/12. In this case, kd equates π/6 or 30 degrees, and so the condition for a directional azimuthal radiation pattern with a null towards phi=0 and maximum gain towards phi=180 is for the current on dipole 204 to lag those on dipole 202 by 150 degrees. At resonance, the currents pass close to this condition (as shown in FIG. 2E), which explains the directionality of the patterns. In the case of the excitation of port 204, the radiation patterns are the mirror opposite of those of FIG. 2F, and maximum gain is in the phi=0 direction. The difference in antenna patterns produced from the two ports has an associated low predicted envelope correlation as shown on FIG. 2G. Thus the combined antenna structure has two ports that are isolated from each other and produce gain patterns of low correlation.

Accordingly, the frequency response of the coupling is dependent on the characteristics of the connecting elements 210, 212, including their impedance and electrical length. In accordance with one or more embodiments of the disclosure, the frequency or bandwidth over which a desired amount of isolation can be maintained is controlled by appropriately configuring the connecting elements. One way to configure the cross connection is to change the physical length of the connecting element. An example of this is shown by the multimode antenna structure 300 of FIG. 3A where a meander has been added to the cross connection path of the connecting elements 310, 312. This has the general effect of increasing both the electrical length and the impedance of the connection between the two antenna elements 302, 304. Performance characteristics of this structure including scattering parameters, current ratios, gain patterns, and pattern correlation are shown on FIGS. 3B, 3C, 3D, and 3E, respectively. In this embodiment, the change in physical length has not significantly altered the resonant frequency of the structure, but there is a significant change in S21, with larger bandwidth and a greater minimum value than in structures without the meander. Thus, it is possible to optimize or improve the isolation performance by altering the electrical characteristic of the connecting elements.

Exemplary multimode antenna structures in accordance with various embodiments of the disclosure can be designed to be excited from a ground or counterpoise 402 (as shown by antenna structure 400 in FIG. 4), or as a balanced structure (as shown by antenna structure 500 in FIG. 5). In either case, each antenna structure includes two or more antenna elements (402, 404 in FIGS. 4, and 502, 504 in FIG. 5) and one or more electrically conductive connecting elements (406 in FIG. 4, and 506, 508 in FIG. 5). For ease of illustration, only a two-port structure is illustrated in the example diagrams. However, it is possible to extend the structure to include more than two ports in accordance with various embodiments of the disclosure. A signal connection to the antenna structure, or port (418, 412 in FIG. 4 and 510, 512 in FIG. 5), is provided at each antenna element. The connecting element provides electrical connection between the two antenna elements at the frequency or frequency range of interest. Although the antenna is physically and electrically one structure, its operation can be explained by considering it as two independent antennas. For antenna structures not including a connecting element such as antenna structure 100, port 106 of that structure can be said to be connected to antenna 102, and port 108 can be said to be connected to antenna 104. However, in the case of this combined structure such as antenna structure 400, port 418 can be referred to as being associated with one antenna mode, and port 412 can be referred to as being associated with another antenna mode.

The antenna elements are designed to be resonant at the desired frequency or frequency range of operation. The lowest order resonance occurs when an antenna element has an electrical length of one quarter of a wavelength. Thus, a simple element design is a quarter-wave monopole in the case of an unbalanced configuration. It is also possible to use higher order modes. For example, a structure formed from quarter-wave monopoles also exhibits dual mode antenna performance with high isolation at a frequency of three times the fundamental frequency. Thus, higher order modes may be exploited to create a multiband antenna. Similarly, in a balanced configuration, the antenna elements can be complementary quarter-wave elements as in a half-wave center-fed dipole. However, the antenna structure can also be formed from other types of antenna elements that are resonant at the desired frequency or frequency range. Other possible antenna element configurations include, but are not limited to, helical coils, wideband planar shapes, chip antennas, meandered shapes, loops, and inductively shunted forms such as Planar Inverted-F Antennas (PIFAs).

The antenna elements of an antenna structure in accordance with one or more embodiments of the disclosure need not have the same geometry or be the same type of antenna element. The antenna elements should each have resonance at the desired frequency or frequency range of operation.

In accordance with one or more embodiments of the disclosure, the antenna elements of an antenna structure have the same geometry. This is generally desirable for design simplicity, especially when the antenna performance requirements are the same for connection to either port.

The bandwidth and resonant frequencies of the combined antenna structure can be controlled by the bandwidth and resonance frequencies of the antenna elements. Thus, broader bandwidth elements can be used to produce a broader bandwidth for the modes of the combined structure as illustrated, e.g., in FIGS. 6A, 6B, and 6C. FIG. 6A illustrates a multimode antenna structure 600 including two dipoles 602, 604 connected by connecting elements 606, 608. The dipoles 602, 604 each have a width (W) and a length (L) and are spaced apart by a distance (d). FIG. 6B illustrates the scattering parameters for the structure having exemplary dimensions: W=1 mm, L=57.2 mm, and d=10 mm. FIG. 6C illustrates the scattering parameters for the structure having exemplary dimensions: W=10 mm, L=50.4 mm, and d=10mm. As shown, increasing W from 1 mm to 10 mm, while keeping the other dimensions generally the same, results in a broader isolation bandwidth and impedance bandwidth for the antenna structure.

It has also been found that increasing the separation between the antenna elements increases the isolation bandwidth and the impedance bandwidth for an antenna structure.

In general, the connecting element is in the high-current region of the combined resonant structure. It may therefore be desirable for the connecting element to have a high conductivity.

The ports are located at the feed points of the antenna elements as they would be if they were operated as separate antennas. Matching elements or structures may be used to match the port impedance to the desired system impedance.

In accordance with one or more embodiments of the disclosure, the multimode antenna structure can be a planar structure incorporated, e.g., into a printed circuit board, as shown as FIG. 7. In this example, the antenna structure 700 includes antenna elements 702, 704 connected by a connecting element 706 at ports 708, 710. The antenna structure is fabricated on a printed circuit board substrate 712. The antenna elements shown in the figure are simple quarter-wave monopoles. However, the antenna elements can be any geometry that yields an equivalent effective electrical length.

In accordance with one or more embodiments of the disclosure, antenna elements with dual resonant frequencies can be used to produce a combined antenna structure with dual resonant frequencies and hence dual operating frequencies. FIG. 8A shows an exemplary model of a multimode dipole structure 800 where the dipole antenna elements 802, 804 are split into two fingers 806, 808 and 810, 812, respectively, of unequal length. The dipole antenna elements have resonant frequencies associated with each the two different finger lengths and accordingly exhibit a dual resonance. Similarly, the multimode antenna structure using dual-resonant dipole arms exhibits two frequency bands where high isolation (or small S21) is obtained as shown in FIG. 8B.

In accordance with one or more embodiments of the disclosure, a multimode antenna structure 900 shown in FIG. 9 is provided having variable length antenna elements 902, 904 forming a tunable antenna. This may be done by changing the effective electrical length of the antenna elements by a controllable device such as an RF switch 906, 908 at each antenna element 902, 904. In this example, the switch may be opened (by operating the controllable device) to create a shorter electrical length (for higher frequency operation) or closed to create a longer electrical length (for lower frequency of operation). The operating frequency band for the antenna structure 900, including the feature of high isolation, can be tuned by tuning both antenna elements in concert. This approach may be used with a variety of methods of changing the effective electrical length of the antenna elements including, e.g., using a controllable dielectric material, loading the antenna elements with a variable capacitor such as a microelectromechanical systems (MEMs) device, varactor, or tunable dielectric capacitor, and switching on or off parasitic elements.

In accordance with one or more embodiments of the disclosure, the connecting element or elements provide an electrical connection between the antenna elements with an electrical length approximately equal to the electrical distance between the elements. Under this condition, and when the connecting elements are attached at the port ends of the antenna elements, the ports are isolated at a frequency near the resonance frequency of the antenna elements. This arrangement can produce nearly perfect isolation at particular frequency.

Alternately, as previously discussed, the electrical length of the connecting element may be increased to expand the bandwidth over which isolation exceeds a particular value. For example, a straight connection between antenna elements may produce a minimum S21 of −25 dB at a particular frequency and the bandwidth for which S21<−10 dB may be 100 MHz. By increasing the electrical length, a new response can be obtained where the minimum S21 is increased to −15 dB but the bandwidth for which S21<−10 dB may be increased to 150 MHz.

Various other multimode antenna structures in accordance with one or more embodiments of the disclosure are possible. For example, the connecting element can have a varied geometry or can be constructed to include components to vary the properties of the antenna structure. These components can include, e.g., passive inductor and capacitor elements, resonator or filter structures, or active components such as phase shifters.

In accordance with one or more embodiments of the disclosure, the position of the connecting element along the length of the antenna elements can be varied to adjust the properties of the antenna structure. The frequency band over which the ports are isolated can be shifted upward in frequency by moving the point of attachment of the connecting element on the antenna elements away from the ports and towards the distal end of the antenna elements. FIGS. 10A and 10B illustrate multimode antenna structures 1000, 1002, respectively, each having a connecting element electrically connected to the antenna elements. In the FIG. 10A antenna structure 1000, the connecting element 1004 is located in the structure such that the gap between the connecting element 1004 and the top edge of the ground plane 1006 is 3 mm. FIG. 10C shows the scattering parameters for the structure showing that high isolation is obtained at a frequency of 1.15 GHz in this configuration. A shunt capacitor/series inductor matching network is used to provide the impedance match at 1.15 GHz. FIG. 10D shows the scattering parameters for the structure 1002 of FIG. 10B, where the gap between the connecting element 1008 and the top edge 1010 of the ground plane is 19 mm. The antenna structure 1002 of FIG. 10B exhibits an operating band with high isolation at approximately 1.50 GHz.

FIG. 11 schematically illustrates a multimode antenna structure 1100 in accordance with one or more further embodiments of the disclosure. The antenna structure 1100 includes two or more connecting elements 1102, 1104, each of which electrically connects the antenna elements 1106, 1108. (For ease of illustration, only two connecting elements are shown in the figure. It should be understood that use of more than two connecting elements is also contemplated.) The connecting elements 1102, 1104 are spaced apart from each other along the antenna elements 1106, 1108. Each of the connecting elements 1102, 1104 includes a switch 1112, 1110. Peak isolation frequencies can be selected by controlling the switches 1110, 1112. For example, a frequency f1 can be selected by closing switch 1110 and opening switch 1112. A different frequency f2 can be selected by closing switch 1112 and opening switch 1110.

FIG. 12 illustrates a multimode antenna structure 1200 in accordance with one or more alternate embodiments of the disclosure. The antenna structure 1200 includes a connecting element 1202 having a filter 1204 operatively coupled thereto. The filter 1204 can be a low pass or band pass filter selected such that the connecting element connection between the antenna elements 1206, 1208 is only effective within the desired frequency band, such as the high isolation resonance frequency. At higher frequencies, the structure will function as two separate antenna elements that are not coupled by the electrically conductive connecting element, which is open circuited.

FIG. 13 illustrates a multimode antenna structure 1300 in accordance with one or more alternate embodiments of the disclosure. The antenna structure 1300 includes two or more connecting elements 1302, 1304, which include filters 1306, 1308, respectively. (For ease of illustration, only two connecting elements are shown in the figure. It should be understood that use of more than two connecting elements is also contemplated.) In one possible embodiment, the antenna structure 1300 has a low pass filter 1308 on the connecting element 1304 (which is closer to the antenna ports) and a high pass filter 1306 on the connecting element 1302 in order to create an antenna structure with two frequency bands of high isolation, i.e., a dual band structure.

FIG. 14 illustrates a multimode antenna structure 1400 in accordance with one or more alternate embodiments of the disclosure. The antenna structure 1400 includes one or more connecting elements 1402 having a tunable element 1406 operatively connected thereto. The antenna structure 1400 also includes antenna elements 1408, 1410. The tunable element 1406 alters the delay or phase of the electrical connection or changes the reactive impedance of the electrical connection. The magnitude of the scattering parameters S21/S12 and a frequency response are affected by the change in electrical delay or impedance and so an antenna structure can be adapted or generally optimized for isolation at specific frequencies using the tunable element 1406.

FIG. 15 illustrates a multimode antenna structure 1500 in accordance with one or more alternate embodiments of the disclosure. The multimode antenna structure 1500 can be used, e.g., in a WIMAX USB dongle. The antenna structure 1500 can be configured for operation, e.g., in WiMAX bands from 2300 to 2700 MHz.

The antenna structure 1500 includes two antenna elements 1502, 1504 connected by a conductive connecting element 1506. The antenna elements include slots to increase the electrical length of the elements to obtain the desired operating frequency range. In this example, the antenna structure is optimized for a center frequency of 2350 MHz. The length of the slots can be reduced to obtain higher center frequencies. The antenna structure is mounted on a printed circuit board assembly 1508. A two-component lumped element match is provided at each antenna feed.

The antenna structure 1500 can be manufactured, e.g., by metal stamping. It can be made, e.g., from 0.2 mm thick copper alloy sheet. The antenna structure 1500 includes a pickup feature 1510 on the connecting element at the center of mass of the structure, which can be used in an automated pick-and-place assembly process. The antenna structure is also compatible with surface-mount reflow assembly.

FIG. 16 illustrates a multimode antenna structure 1600 in accordance with one or more alternate embodiments of the disclosure. As with antenna structure 1500 of FIG. 15, the antenna structure 1600 can also be used, e.g., in a WIMAX USB dongle. The antenna structure can be configured for operation, e.g., in WiMAX bands from 2300 to 2700 MHz.

The antenna structure 1600 includes two antenna elements 1602, 1604, each comprising a meandered monopole. The length of the meander determines the center frequency. The exemplary design shown in the figure is optimized for a center frequency of 2350 MHz. To obtain higher center frequencies, the length of the meander can be reduced.

A connecting element 1606 electrically connects the antenna elements. A two-component lumped element match is provided at each antenna feed.

The antenna structure can be fabricated, e.g., from copper as a flexible printed circuit (FPC) mounted on a plastic carrier 1608. The antenna structure can be created by the metalized portions of the FPC. The plastic carrier provides mechanical support and facilitates mounting to a PCB assembly 1610. Alternatively, the antenna structure can be formed from sheet-metal.

FIG. 17 illustrates a multimode antenna structure 1700 in accordance with another embodiment of the disclosure. This antenna design can be used, e.g., for USB, Express 34, and Express 54 data card formats. The exemplary antenna structure shown in the figure is designed to operate at frequencies from 2.3 to 6 GHz. The antenna structure can be fabricated, e.g., from sheet-metal or by FPC over a plastic carrier 1702.

FIG. 18A illustrates a multimode antenna structure 1800 in accordance with another embodiment of the disclosure. The antenna structure 1800 comprises a three mode antenna with three ports. In this structure, three monopole antenna elements 1802, 1804, 1806 are connected using a connecting element 1808 comprising a conductive ring that connects neighboring antenna elements. The antenna elements are balanced by a common counterpoise, or sleeve 1810, which is a single hollow conductive cylinder. The antenna has three coaxial cables 1812, 1814, 1816 for connection of the antenna structure to a communications device. The coaxial cables 1812, 1814, 1816 pass through the hollow interior of the sleeve 1810. The antenna assembly may be constructed from a single flexible printed circuit wrapped into a cylinder and may be packaged in a cylindrical plastic enclosure to provide a single antenna assembly that takes the place of three separate antennas. In one exemplary arrangement, the diameter of the cylinder is 10 mm and the overall length of the antenna is 56 mm so as to operate with high isolation between ports at 2.45 GHz. This antenna structure can be used, e.g., with multiple antenna radio systems such as MIMO or 802.11N systems operating in the 2.4 to 2.5 GHz bands. In addition to port to port isolation, each port advantageously produces a different gain pattern as shown on FIG. 18B. While this is one specific example, it is understood that this structure can be scaled to operate at any desired frequency. It is also understood that methods for tuning, manipulating bandwidth, and creating multiband structures described previously in the context of two-port antennas can also apply to this multiport structure.

While the above embodiment is shown as a true cylinder, it is possible to use other arrangements of three antenna elements and connecting elements that produce the same advantages. This includes, but is not limited to, arrangements with straight connections such that the connecting elements form a triangle, or another polygonal geometry. It is also possible to construct a similar structure by similarly connecting three separate dipole elements instead of three monopole elements with a common counterpoise. Also, while symmetric arrangement of antenna elements advantageously produces equivalent performance from each port, e.g., same bandwidth, isolation, impedance matching, it is also possible to arrange the antenna elements asymmetrically or with unequal spacing depending on the application.

FIG. 19 illustrates use of a multimode antenna structure 1900 in a combiner application in accordance with one or more embodiments of the disclosure. As shown in the figure, transmit signals may be applied to both antenna ports of the antenna structure 1900 simultaneously. In this configuration, the multimode antenna can serve as both antenna and power amplifier combiner. The high isolation between antenna ports restricts interaction between the two amplifiers 1902, 1904, which is known to have undesirable effects such as signal distortion and loss of efficiency. Optional impedance matching at 1906 can be provided at the antenna ports.

FIGS. 20A and 20B illustrate a multimode antenna structure 2000 in accordance with one or more alternate embodiments of the subject disclosure. The antenna structure 2000 can also be used, e.g., in a WiMAX USB or ExpressCard/34 device. The antenna structure can be configured for operation, e.g., in WiMAX bands from 2300 to 6000 MHz.

The antenna structure 2000 includes two antenna elements 2001, 2004, each comprising a broad monopole. A connecting element 2002 electrically connects the antenna elements. Slots (or other cut-outs) 2005 are used to improve the input impedance match above 5000 MHz. The exemplary design shown in the figure is optimized to cover frequencies from 2300 to 6000 MHz.

The antenna structure 2000 can be manufactured, e.g., by metal stamping. It can be made, e.g., from 0.2 mm thick copper alloy sheet. The antenna structure 2000 includes a pickup feature 2003 on the connecting element 2002 generally at the center of mass of the structure, which can be used in an automated pick-and-place assembly process. The antenna structure is also compatible with surface-mount reflow assembly. Feed points 2006 of the antenna provide the points of connection to the radio circuitry on a PCB, and also serve as a support for structural mounting of the antenna to the PCB. Additional contact points 2007 provide structural support.

FIG. 20C illustrates a test assembly 2010 used to measure the performance of antenna 2000. The figure also shows the coordinate reference for far-field patterns. Antenna 2000 is mounted on a 30×88 mm PCB 2011 representing an ExpressCard/34 device. The grounded portion of the PCB 2011 is attached to a larger metal sheet 2012 (having dimensions of 165×254 mm in this example) to represent a counterpoise size typical of a notebook computer. Test ports 2014, 2016 on the PCB 2011 are connected to the antenna through 50-ohm striplines.

FIG. 20D shows the VSWR measured at test ports 2014, 2016. FIG. 20E shows the coupling (S21 or S12) measured between the test ports. The VSWR and coupling are advantageously low across the broad range of frequencies, e.g., 2300 to 6000 MHz. FIG. 20F shows the measured radiation efficiency referenced from the test ports 2014 (Port 1), 2016 (Port 2). FIG. 20G shows the calculated correlation between the radiation patterns produced by excitation of test port 2014 (Port 1) versus those produced by excitation of test port 2016 (Port 2). The radiation efficiency is advantageously high while the correlation between patterns is advantageously low at the frequencies of interest. FIG. 20H shows far field gain patterns by excitation of test port 2014 (Port 1) or test port 2016 (Port 2) at a frequency of 2500 MHz. FIGS. 20I and 20J show the same pattern measurements at frequencies of 3500 and 5200 MHz, respectively. The patterns resulting from test port 2014 (Port 1) are different and complementary to those of test port 2016 (Port 2) in the φ=0 or XZ plane and in the θ=90 or XY plane.

FIGS. 21A and 21B illustrate a multimode antenna structure 2100 in accordance with one or more alternate embodiments of the subject disclosure. The antenna structure 2100 can also be used, e.g., in a WiMAX USB dongle. The antenna structure can be configured for operation, e.g., in WiMAX bands from 2300 to 2400 MHz.

The antenna structure 2100 includes two antenna elements 2102, 2104, each comprising a meandered monopole. The length of the meander determines the center frequency. Other tortuous configurations such as, e.g., helical coils and loops, can also be used to provide a desired electrical length. The exemplary design shown in the figure is optimized for a center frequency of 2350 MHz. A connecting element 2106 (shown in FIG. 21B) electrically connects the antenna elements 2102, 2104. A two-component lumped element match is provided at each antenna feed.

The antenna structure can be fabricated, e.g., from copper as a flexible printed circuit (FPC) 2103 mounted on a plastic carrier 2101. The antenna structure can be created by the metalized portions of the FPC 2103. The plastic carrier 2101 provides mounting pins or pips 2107 for attaching the antenna to a PCB assembly (not shown) and pips 2105 for securing the FPC 2103 to the carrier 2101. The metalized portion of 2103 includes exposed portions or pads 2108 for electrically contacting the antenna to the circuitry on the PCB.

To obtain higher center frequencies, the electrical length of the elements 2102, 2104 can be reduced. FIGS. 22A and 22B illustrate a multimode antenna structure 2200, the design of which is optimized for a center frequency of 2600 MHz. The electrical length of the elements 2202, 2204 is shorter than that of elements 2102, 2104 of FIGS. 21A and 21B because metallization at the end of the elements 2202, 2204 has been removed, and the width of the of the elements at feed end has been increased.

FIG. 23A illustrates a test assembly 2300 using antenna 2100 of FIGS. 21A and 21B along with the coordinate reference for far-field patterns. FIG. 23B shows the VSWR measured at test ports 2302 (Port 1), 2304 (Port 2). FIG. 23C shows the coupling (S21 or S12) measured between the test ports 2302 (Port 1), 2304 (Port 2). The VSWR and coupling are advantageously low at the frequencies of interest, e.g.,2300 to 2400 MHz. FIG. 23D shows the measured radiation efficiency referenced from the test ports. FIG. 23E shows the calculated correlation between the radiation patterns produced by excitation of test port 2302 (Port 1) versus those produced by excitation of test port 2304 (Port 2). The radiation efficiency is advantageously high while the correlation between patterns is advantageously low at the frequencies of interest. FIG. 23F shows far field gain patterns by excitation of test port 2302 (Port 1) or test port 2304 (Port 2) at a frequency of 2400 MHz. The patterns resulting from test port 2302 (Port 1) are different and complementary to those of test port 2304 (Port 2) in the φ=0 or XZ plane and in the θ=90 or XY plane.

FIG. 23G shows the VSWR measured at the test ports of assembly 2300 with antenna 2200 in place of antenna 2100. FIG. 23H shows the coupling (S21 or S12) measured between the test ports. The VSWR and coupling are advantageously low at the frequencies of interest, e.g. 2500 to 2700 MHz. FIG. 23I shows the measured radiation efficiency referenced from the test ports. FIG. 23J shows the calculated correlation between the radiation patterns produced by excitation of test port 2302 (Port 1) versus those produced by excitation of test port 2304 (Port 2). The radiation efficiency is advantageously high while the correlation between patterns is advantageously low at the frequencies of interest. FIG. 23K shows far field gain patterns by excitation of test port 2302 (Port 1) or test port 2304 (Port 2) at a frequency of 2600 MHz. The patterns resulting from test port 2302 (Port 1) are different and complementary to those of test port 2304 (Port 2) in the φ=0 or XZ plane and in the θ=90 or XY plane.

One or more further embodiments of the subject disclosure are directed to techniques for beam pattern control for the purpose of null steering or beam pointing. When such techniques are applied to a conventional array antenna (comprising separate antenna elements that are spaced at some fraction of a wavelength), each element of the array antenna is fed with a signal that is a phase shifted version of a reference signal or waveform. For a uniform linear array with equal excitation, the beam pattern produced can be described by the array factor F, which depends on the phase of each individual element and the inter-element element spacing d.

$F = {A_{0}{\sum\limits_{n = 0}^{N - 1}\; {\exp \left\lbrack {j\; {n\left( {{\beta \; d\mspace{14mu} \cos \mspace{14mu} \theta} + \alpha} \right)}} \right\rbrack}}}$

where β=2π/λ, N=Total # of elements, α=phase shift between successive elements, and θ=angle from array axis

By controlling the phase α to a value α_(i), the maximum value of F can be adjusted to a different direction θ_(i), thereby controlling the direction in which a maximum signal is broadcast or received.

The inter-element spacing in conventional array antennas is often on the order of ¼ wavelength, and the antennas can be closely coupled, having nearly identical polarization. It is advantageous to reduce the coupling between elements, as coupling can lead to several problems in the design and performance of array antennas. For example, problems such as pattern distortion and scan blindness (see Stutzman, Antenna Theory and Design, Wiley 1998, pp. 122-128 and 135-136, and 466-472) can arise from excessive inter-element coupling, as well as a reduction of the maximum gain attainable for a given number of elements.

Beam pattern control techniques can be advantageously applied to all multimode antenna structures described herein having antenna elements connected by one or more connecting elements, which exhibit high isolation between multiple feedpoints. The phase between ports at the high isolation antenna structure can be used for controlling the antenna pattern. It has been found that a higher peak gain is achievable in given directions when the antenna is used as a simple beam-forming array as a result of the reduced coupling between feedpoints. Accordingly, greater gain can be achieved in selected directions from a high isolation antenna structure in accordance with various embodiments that utilizes phase control of the carrier signals presented to its feed terminals.

In handset applications where the antennas are spaced at much less than ¼ wavelength, mutual coupling effects in conventional antennas reduce the radiation efficiency of the array, and therefore reduce the maximum gain achievable.

By controlling the phase of the carrier signal provided to each feedpoint of a high isolation antenna in accordance with various embodiments, the direction of maximum gain produced by the antenna pattern can be controlled. A gain advantage of, e.g., 3 dB obtained by beam steering is advantageous particularly in portable device applications where the beam pattern is fixed and the device orientation is randomly controlled by the user. As shown, e.g., in the schematic block diagram of FIG. 24, which illustrates a pattern control apparatus 2400 in accordance with various embodiments, a relative phase shift cc is applied by a phase shifter 2402 to the RF signals applied to each antenna feed 2404, 2408. The signals are fed to respective antenna ports of antenna structure 2410.

The phase shifter 2402 can comprise standard phase shift components such as, e.g., electrically controlled phase shift devices or standard phase shift networks.

FIGS. 25A-25G provide a comparison of antenna patterns produced by a closely spaced 2-D conventional array of dipole antennas and a 2-D array of high isolation antennas in accordance with various embodiments of the subject disclosure for different phase differences cc between two feeds to the antennas. In FIGS. 25A-25G, curves are shown for the antenna patterns at θ=90 degrees. The solid lines in the figures represents the antenna pattern produced by the isolated feed single element antenna in accordance with various embodiments, while the dashed lines represent the antenna pattern produced by two separate monopole conventional antennas separated by a distance equal to the width of the single element isolated feed structure. Therefore, the conventional antenna and the high isolation antenna are of generally equivalent size.

In all cases shown in the figures, the peak gain produced by the high isolation antenna in accordance with various embodiments produces a greater gain margin when compared to the two separate conventional dipoles, while providing azimuthal control of the beam pattern. This behavior makes it possible to use the high isolation antenna in transmit or receive applications where additional gain is needed or desired in a particular direction. The direction can be controlled by adjusting the relative phase between the drivepoint signals. This may be particularly advantageous for portable devices needing to direct energy toward a receive point such as, e.g., a base station. The combined high isolation antenna offers greater advantage when compared to two single conventional antenna elements when phased in a similar fashion.

As shown in FIG. 25A, the combined dipole in accordance with various embodiments shows greater gain in a uniform azimuth pattern (θ=90) for α=0 (zero degrees phase difference).

As shown in FIG. 25B, the combined dipole in accordance with various embodiments shows greater peak gain (at φ=0) with a non-symmetric azimuthal pattern (θ=90 plot for α=30 (30 degrees phase difference between feedpoints).

As shown in FIG. 25C, the combined dipole in accordance with various embodiments shows greater peak gain (at φ=0) with a shifted azimuthal pattern (θ=90 plot for α=60 (60 degrees phase difference between feedpoints).

As shown in FIG. 25D, the combined dipole in accordance with various embodiments shows even greater peak gain (at φ=0) with a shifted azimuthal pattern (θ=90 plot for α=90 (90 degrees phase difference between feedpoints).

As shown in FIG. 25E, the combined dipole in accordance with various embodiments shows greater peak gain (at φ=0) with a shifted azimuthal pattern (θ=90 plot greater backlobe (at φ=180) for α=120 (120 degrees phase difference between feedpoints).

As shown in FIG. 25F, the combined dipole in accordance with various embodiments shows greater peak gain (at φ=0) with a shifted azimuthal pattern (θ=90 plot), even greater backlobe (at φ=180) for α=150 (150 degrees phase difference between feedpoints).

As shown in FIG. 25G, the combined dipole in accordance with various embodiments shows greater peak gain (at φ=0 & 180) with a double lobed azimuthal pattern (θ=90 plot) for α=180 (180 degrees phase difference between feedpoints).

FIG. 26 illustrates the ideal gain advantage if the combined high isolation antenna in accordance with one or more embodiments over two separate dipoles as a function of the phase angle difference between the feedpoints for a two feedpoint antenna array.

Further embodiments of the subject disclosure are directed to multimode antenna structures that provide increased high isolation between multi-band antenna ports operating in close proximity to each other at a given frequency range. In these embodiments, a band-rejection slot is incorporated in one of the antenna elements of the antenna structure to provide reduced coupling at the frequency to which the slot is tuned.

FIG. 27A schematically illustrates a simple dual-band branch line monopole antenna 2700. The antenna 2700 includes a band-rejection slot 2702, which defines two branch resonators 2704, 2706. The antenna is driven by signal generator 2708. Depending on the frequency at which the antenna 2700 is driven, various current distributions are realized on the two branch resonators 2704, 2706.

The physical dimensions of the slot 2702 are defined by the width Ws and the length Ls as shown in FIG. 27A. When the excitation frequency satisfies the condition of Ls=lo/4, the slot feature becomes resonant. At this point the current distribution is concentrated around the shorted section of the slot, as shown in FIG. 27B.

The currents flowing through the branch resonators 2704, 2706 are approximately equal and oppositely directed along the sides of the slot 2702. This causes the antenna structure 2700 to behave in a similar manner to a spurline band stop filter 2720 (shown schematically in FIG. 27C), which transforms the antenna input impedance down significantly lower than the nominal source impedance. This large impedance mismatch results in a very high VSWR, shown in FIGS. 27D and 27E, and as a result leads to the desired frequency rejection.

This band-rejection slot technique can be applied to an antenna system with two (or more) antennas elements operating in close proximity to each other where one antenna element needs to pass signals of a desired frequency and the other does not. In one or more embodiments, one of the two antenna elements includes a band-rejection slot, and the other does not. FIG. 28 schematically illustrates an antenna structure 2800, which includes a first antenna element 2802, a second antenna element 2804, and a connecting element 2806. The antenna structure 2800 includes ports 2808 and 2810 at antenna elements 2802 and 2804, respectively. In this example, a signal generator drives the antenna structure 2802 at port 2808, while a meter is coupled to the port 2810 to measure current at port 2810. It should be understood, however, that either or both ports can be driven by signal generators. The antenna element 2802 includes a band-rejection slot 2812, which defines two branch resonators 2814, 2816. In this embodiment, the branch resonators comprise the main transmit section of the antenna structure, while the antenna element 2804 comprises a diversity receive portion of the antenna structure.

Due to the large mismatch at the port of the antenna element 2802 with the band-reject slot 2812, the mutual coupling between it and the diversity receive antenna element 2804, which is actually matched at the slot resonant frequency will be quite small and will result in relatively high isolation.

FIG. 29A is a perspective view of a multimode antenna structure 2900 comprising a multi-band diversity receive antenna system that utilizes the band-rejection slot technique in the GPS band in accordance with one or more further embodiments of the subject disclosure. (The GPS band is 1575.42 MHz with 20 MHz bandwidth.) The antenna structure 2900 is formed on a flex film dielectric substrate 2902, which is formed as a layer on a dielectric carrier 2904. The antenna structure 2900 includes a GPS band rejection slot 2906 on the primary transmit antenna element 2908 of the antenna structure 2900. The antenna structure 2900 also includes a diversity receive antenna element 2910, and a connecting element 2912 connecting the diversity receive antenna element 2910 and the primary transmit antenna element 2908. A GPS receiver (not shown) is connected to the diversity receive antenna element 2910. In order to generally minimize the antenna coupling from the primary transmit antenna element 2908 and to generally maximize the diversity antenna radiation efficiency at these frequencies, the primary antenna element 2908 includes the band-rejection slot 2906 and is tuned to an electrical quarter wave length near the center of the GPS band. The diversity receive antenna element 2910 does not contain such a band rejection slot, but comprises a GPS antenna element that is properly matched to the main antenna source impedance so that there will be generally maximum power transfer between it and the GPS receiver. Although both antenna elements 2908, 2910 co-exist in close proximity, the high VSWR due to the slot 2906 at the primary transmit antenna element 2908 reduces the coupling to the primary antenna element source resistance at the frequency to which the slot 2906 is tuned, and therefore provides isolation at the GPS frequency between both antenna elements 2908, 2910. The resultant mismatch between the two antenna elements 2908, 2910 within the GPS band is large enough to decouple the antenna elements in order to meet the isolation requirements for the system design as shown in FIGS. 29B and 29C.

In the antenna structures described herein in accordance with various embodiments of the subject disclosure, the antenna elements and the connecting elements can form a single integrated radiating structure such that a signal fed to either port excites the entire antenna structure to radiate as a whole, rather than separate radiating structures. As such, the techniques described herein provide isolation of the antenna ports without the use of decoupling networks at the antenna feed points.

Other embodiments disclosed herein are directed to an antenna that separates the fundamental (low band) resonance from the high band resonance by using two separate structures, which are connected at the feedpoint—thus accomplishing the goal of achieving a MIMO or Diversity antenna with each feed exhibiting a multiband capability, and whereby each feed is optimally isolated from the opposite feed. By way of a non-limiting illustration, in some implementations, high band frequencies can range from 1710 to 2170 MHz, and low band frequencies can range from 698 to 960 MHz.

In one or more embodiments of the antenna structures described in the subject disclosure, electrical currents flowing through neighboring antenna elements 3002 and 3004 (see FIG. 30) can be configured to be similar in magnitude, such that an antenna mode excited by one antenna port (e.g., Port 1) is approximately electrically isolated from an antenna mode excited by another antenna port (e.g., Port 2) at a given desired signal frequency range. In one embodiment, this can be accomplished by configuring antennas 3002 and 3004 with a connecting element 3006 to enable common and difference mode currents, which when summed together result in some or a substantial amount of isolation between ports 1 and 2. Configuring an antenna structure to control differential and common mode current to cause isolation between any number of antenna ports can be applied to any of the antenna embodiments described herein.

FIG. 31 illustrates an exemplary multiband antenna 3100 in accordance with one or more embodiments. The antenna 3100 can include a low band structure comprising two low band antenna elements 3102, 3104 connected by a connecting element 3106. A fixed or variable reactive element 3126 such as a fixed or variable inductor L is provided in the connecting element 3106 to provide control (reduction) of the mutual coupling between feedpoints for the low band element by varying the electrical length of the connecting element 3106 in accordance with the disclosures of U.S. Pat. No. 7,688,273, the disclosure of which is incorporated by reference herein in its entirety. Similarly, a connecting element 3116 can be provided between the high band antenna elements 3112, 3114. A fixed or variable reactive element 3136 such as a fixed or variable inductor L can be provided in the connecting element 3116 to provide control (reduction) of the mutual coupling between feedpoints for the low band element by varying the electrical length of the connecting element 3116 in accordance with the disclosures of U.S. Pat. No. 7,688,273.

The high band structure comprising two high band antenna elements 3112, 3114 can be connected to the low band structure at feed points f1, f2. Two filters 3142 and 3144 are provided in the high band antenna elements 3112, 3114 for blocking low band frequencies, thereby isolating the high band antenna elements 3112, 3114 from the low band antenna elements 3102, 3104. The filters 3142 and 3144 can be passive or programmable pass band filters. In the present illustration the filters 3142 and 3144 can represent high pass filters implemented with a capacitor and/or other components to achieve desired high pass filtering characteristics. To achieve similar isolation with the low band structure, the low band antenna elements 3102, 3104 can be configured with filters 3152, 3154 to block high band frequencies, thereby isolating the high band antenna elements 3112, 3114 from the low band antenna elements 3102, 3104. The filters 3152, 3154 can be passive or programmable pass band filters. In the present illustration the filters 3152, 3154 can represent low pass filters implemented with reactive and passive components that achieve desired low pass filtering characteristics.

By having a structure associated with low band resonance and a separate structure associated with high band resonance, the low band structure can be advantageously designed or optimized independently of the high band structure and vice-versa. A further advantage is that the low band or high band structures may separately take on different antenna design realizations, e.g., monopole, loop, Planar Inverted “F” antenna (PIFA), etc. allowing the designer to select the best option for the electrical and mechanical design requirements. In one exemplary embodiment, the low band structure may be a monopole, while the high band structure may be a PIFA.

A separate network is provided for each structure. The low band structure can use a fixed or variable inductive bridge 3126 as an interconnecting element 3106. The high band element is fed from the common feedpoint, but with a high pass network 3142, 3144—the simplest being a series capacitor with low reactance at the high band frequencies and higher reactance at the low band frequencies. In addition, the low band antenna elements 3102, 3104 can be configured with variable reactive components 3122, 3124 to perform aperture tuning which enables shifting of the low band resonance frequency of the low band structure. The reactive components 3122, 3124 can be independently controlled so that the resonance frequency of low band antenna element 3102 can be independently controlled from the low band resonance frequency of low band antenna element 3104. The reactive components 3122, 3124 can be represented by switched inductors which can be aggregated or reduced to vary the electrical length of the low band antenna elements 3102, 3104, respectively.

Similarly, the high band antenna elements 3112, 3114 can be configured with variable reactive components 3132, 3134 to perform aperture tuning which enables shifting of the high band resonance frequency of the high band structure. The reactive components 3132, 3134 can be independently controlled so that the resonance frequency of high band antenna element 3112 can be independently controlled from the high band resonance frequency of high band antenna element 3114. The reactive components 3132, 3134 can also be represented by switched inductors which can be aggregated or reduced to vary the electrical length of the high band antenna elements 3112, 3114, respectively.

The aforementioned structures, enable high band tuning to be performed relatively independent of low band tuning, providing a simpler design process and better performance than antennas not having such separate structures. Other more complex networks may also be used advantageously to separate the interdependence of the high and low band structures still using a common feedpoint for a MIMO branch such as shown in FIG. 31. The method illustrated in FIG. 31 is not limited to 2×2, 2×1 MIMO or 2 feed antennas used for diversity applications, and may be extended to higher branch order MIMO antennas, e.g., 3×3, etc.

A number of factors affect antenna performance in a hand held mobile communication device. While these factors are related, they generally fall into one of three categories; antenna size, mutual coupling between multiple antennas, and device usage models. The size of an antenna is dependent on three criteria; bandwidth of operation, frequency of operation, and required radiation efficiency. Bandwidth requirements have obviously increased as they are driven by FCC frequency allocations in the US and carrier roaming agreements around the world. Different regions use different frequency bands, now with over 40 E-UTRA band designations-many overlapping but requiring world capable wireless devices to typically cover a frequency range from 698 to 2700 MHz.

A simple relationship exists between the bandwidth, size, and radiation efficiency for the fundamental or lowest frequency resonance of a physically small antenna.

$\begin{matrix} {\frac{\Delta \; f}{f} \propto {\left( \frac{a}{\lambda} \right)^{3}\eta^{- 1}}} & (1) \end{matrix}$

Here a is the radius of a sphere containing the antenna and its associated current distribution. Since a is normalized to the operating wavelength, the formula may be interpreted as “fractional bandwidth is proportional to the wavelength normalized modal volume”. The radiation efficiency η is included as a factor on the right side of (1), indicating that greater bandwidth, is achievable by reducing the efficiency. Radio frequency currents exist not only on the antenna element but also on the attached conductive structure or “counterpoise”. For instance, mobile phone antennas in the 698-960 MHz bands use the entire PCB as a radiating structure so that the physical size of the antenna according to (1) is actually much larger than what appears to be the “antenna”. The “antenna” may be considered a resonator that is electromagnetically coupled to the PCB so that it excites currents over the entire conductive structure or chassis. Most smartphones exhibit conductive chassis dimensions of approximately 70×130 mm, which from an electromagnetic modal analysis predicts a fundamental mode near 1 GHz suggesting that performance bandwidth degrades progressively at lower excitation frequencies. The efficiency-bandwidth trade-off is complex requiring E-M simulation tools for accurate prediction. Results indicate that covering 698-960 MHz (Bands 12, 13, 17, 18, 19, 20, 5 and 8) with a completely passive antenna with desirable antenna size and geometry becomes difficult without making sacrifices in radiation efficiency.

Factors determining the achievable radiation efficiency are not entirely obvious, as the coupling coefficient between the “antenna” and the chassis; radiative coupling to lossy components on the PCB; dielectric absorption in plastic housing, coupling to co-existing antennas; as well as losses from finite resistance within the “antenna” resonator structure, all play a part. In most cases, the requirements imposed by operators suggest minimum radiation efficiencies of 40-50%, so that meeting a minimum TRP requirement essentially requires tradeoffs between the power amplifier (PA) output and the achievable antenna efficiency. In turn, poor efficiency at the antenna translates to less battery life, as the PA must compensate for the loss.

Prior to concerns over band aggregation, wireless devices operated on one band at a time with need to change when roaming. Consequently, the required instantaneous bandwidth would be considerably less than that required to address worldwide compatibility. Take a 3G example for instance, where operation in band 5 from (824-894 MHz) compared to operation in bands 5 plus 8 (824-960 MHz). Then, add the requirements for band 13 and band 17 and the comparison becomes more dramatic—824-960 vs. 698-960 MHz. This becomes a problematic as legacy phone antennas support pentaband operation but only bands 5 and band 8. Given equation (1) several choices exist. The most obvious would be to increase the antenna system size, (i.e. the antenna and phone chassis footprint) and/or to reduce the radiation efficiency. Since 4G smartphones require 2 antennas, neither approach is necessarily desirable from an industrial design standpoint, although it is possible to cover the 700-2200 MHz bands with a completely passive antenna in a space allocation of 6.5×10×60 mm.

Various alternative antenna configurations are the following: limit the antenna(s) instantaneous bandwidth within current antenna space allocations to allow use of 1 or more antennas without compromising the industrial design (Antenna Supplier motivation); make the antenna(s) smaller to achieve a compact and sleek device with greater functionality by limiting the instantaneous bandwidth with same or improved antenna efficiency (OEM motivation); improve the antenna efficiency, and therefore the network performance by controlling the antenna instantaneous frequency/tuning (Operator motivation); make the antenna agile to adapt to different usage models (OEM/User/Operator motivation); or combinations of the above.

The simplest approach can be to limit the instantaneous operation to a single band to satisfy the protocol requirements for a single region. To satisfy the roaming requirements, the antenna could be made frequency agile on a band-by-band basis. This approach represents the most basic type of “state-tuned” antenna.

Various embodiments disclosed herein are directed to an antenna that separates the fundamental (low band) resonance from the high band resonance by using two separate structures, which are connected at the feedpoint—thus accomplishing the goal of achieving a MIMO or Diversity antenna with each feed exhibiting a multiband capability, and whereby each feed is optimally isolated from the opposite feed. By way of non-limiting example, in some implementations, high band frequencies can range from 1710 to 2700 MHz, and low band frequencies can range from 500 to 960 MHz.

The exemplary embodiments allow for tuning of the first resonance of the antenna to accommodate multiple operational bands depending on a tuning state, and broadband operation on the high bands (e.g., 1710-2170 MHz, or 1710-2700 MHz) independent of the low band tuning state.

Referring to FIG. 32A, an example is shown that is illustrative of single low band-multiple high band aggregation compatibility. The high band radiation efficiency in this case can remain essentially the same independent of the low band tuning state, but the low band resonance frequency is able to be tuned in discrete frequency increments according to the equivalent electrical length, as selected by the series inductance Lvar which is shown in FIG. 32B. The variable inductance can be created using discrete reactive elements such as inductors and a switching mechanism such as an SP4T switch. The configuration as shown yields 3 different inductances depending on which state the switch is in: (state 1) LVAR=L3+L4+L5 (switch connects to pole 1 or 4); (state 2) LVAR=Lpath2∥L3+L4+L5 or approximately L4+L5 (switch connects to pole 2); or (state 3) LVAR=Lpath3∥(L3+L4)+L5 OR approximately L5 (switch connects to pole 3). In this embodiment, Lpath2 and Lpath3 refer to the equivalent inductances of the circuit paths through the switch. Keeping the inductors close to the switch can minimize or otherwise reduce the path inductances such that the discrete inductors are essentially shorted out by the switch.

The antenna incorporates a main structure that has a fundamental resonance at the lowest frequency band. The solution employs a multiband antenna having 3 low band tuning states as shown in FIG. 32B. State 1 includes a low band (fundamental) resonance suited for LTE 700 (698-742 MHz) operation: State 2 includes a low band resonance suited to GSM 850 (824-894 MHz) operation, and state 3 a low band resonance suited to GSM 900 (880-960 MHz).

The high band resonance (1710-2170 MHz) can be reasonably independent of the tuning state for the low band by nature of the separation of the low and high band radiating elements from the feedpoints. The low band tuning can be accomplished by switching different reactive components in between the feedpoint and the radiating structure. The high band operation of the antenna can be governed primarily by the auxiliary radiating section at the terminus of the capacitor opposite the feedpoint. The capacitor functions primarily as a high pass filter to decouple the feedpoint from the high and low bands portions of the antenna. In this way, signals at different operating bands can be directed to the appropriate radiating section of the combined antenna. The high band resonance can be determined in part by the electrical length of the high band portion of the antenna (indicated in the illustration by horizontal conductive segments). In other embodiments, the capacitor may be a highpass, bandpass, or tunable filter. In a similar manner, the path from the feedpoint to the low band radiating portion of the antenna may include a low pass, bandpass or tunable filter.

Tuning can be accomplished using a switching device such one capable of SP4T operation. In one embodiment, a solid state silicon-based FET switch can be used in each leg of the antenna to alter the series inductance presented to the antenna feedpoint, thereby lowering the resonant frequency as a function of the amount of inductance added. Although inductors are used in this embodiment, other reactive components may also be used for the purpose of altering the electrical length of the low band portion of the antenna radiating structure including capacitive elements. The switch may be of various types such as a mechanical MEMS type device, a voltage/current controlled variable device, and so forth. The switch may also be configured with multiple poles and with any throw capability needed to select the number of tuning states required for antenna operation. The number of throws can establish the number of tuning states possible, which in turn is dictated by the number of frequency bands to be supported. While three states are shown in the illustrated embodiment, any number of states can be utilized corresponding to any number of frequency bands or ranges. In one embodiment, a pair of adjustable reactive elements (e.g., fixed inductors coupled with switching mechanisms) can be coupled with corresponding pairs of feedpoints, and the tuning can be performed by settings each of the adjustable reactive elements to the same tuning state among the group of tuning states.

Referring to FIG. 33A, a separate but co-located high band element is shown with feed points F1H and F2H that allows for compatibility with RF transceiver front end designs requiring separate low- and mid- or low- and high- band connections to the antenna. The variable inductance can be created using discrete inductors and a SP4T switch as shown. The configuration as shown yields 3 different inductances depending on which state the switch is in: (state 1) LVAR=L3+L4+L5 (switch connects to pole 1 or 4); (state 2) LVAR=Lpath2∥L3+L4+L5 or approximately L4+L5 (switch connects to pole 2); or (state 3) LVAR=Lpath3∥(L3+L4)+L5 OR approx. L5 (switch connects to pole 3). Lpath2 and Lpath3 refer to the equivalent inductances of the circuit paths through the switch. Keeping the inductors close to the switch minimizes the path inductances such that the discrete inductors are essentially shorted out by the switch.

The exemplary antennas can provide better radiation efficiency and/or smaller size compared to an untuned antenna by nature of the tuning to each band of operation separately. The reactive elements (e.g., inductors and their associated inductance) can establish the electrical length of the low band elements, and therefore can provide for adjusting the low band resonance (tuning). Referring additionally to FIGS. 34A, 34B, 35A and 35B, antenna structures that enable tuning to each band of operation separately while also providing for desired filtering through use of low-pass-filters and high-pass-filters as illustrated.

Further, the fundamental mode associated of the antenna low band resonance can be tuned by adjustment of the electrical length of the low band portion of the antenna via reactive elements which may exhibit either inductive or capacitive characteristics. As illustrated in FIG. 32A, discrete inductors are shown in a series connection between the antenna feed points and the radiating element end plates on the each side of the antenna, thereby increasing the equivalent electrical length. The use of separate or discrete components is intended to be illustrative of the principle, but by no means limiting to scope of the subject disclosure. In one or more embodiments, the techniques and/or components of the exemplary embodiments described herein that provide for antenna tuning can be utilized in conjunction with techniques and/or components described with respect to U.S. Pat. No. 7,688,273.

A plurality of antennas can be included in a communication device, thereby providing an opportunity to select a combination of antennas with selective polarization. In particular, a set of antennas can be selected to reduce the envelope correlation coefficient (ECC) of the antenna configuration of the device, and/or increase antenna isolation in the device. As is understood in the art, the envelope correlation coefficient p of two antennas is related to the S parameters of the antennas by

$\rho = \frac{{{{S_{11}S_{12}} + {S_{21}S_{22}}}}^{2}}{\left( {1 - {S_{11}}^{2} - {S_{21}}^{2}} \right)\left( {1 - {S_{22}}^{2} - {S_{12}}^{2}} \right)}$

FIG. 36 depicts an illustrative embodiment of polarization selectivity in a multi-antenna device 3600 having two antennas 3610, 3620. As shown in FIG. 36, both antennas are connected to a ground plane 3601 (for example, a conducting material on a printed circuit board as shown in FIG. 7). In this embodiment, antennas 3610, 3620 are both in the x-y plane and have electric field vectors E pointing toward each other. The orientation of two antennas at 180° relative to each other effectively reduces the ECC.

FIG. 37 depicts an illustrative embodiment of polarization selectivity in a multi-antenna device 3700 having two antennas in different planes. In this embodiment, antennas 3710 and 3720 are oriented at 180° relative to each other and are also in different planes; antenna 3710 is in the x-y plane while antenna 3720 is in the y-z plane. This antenna configuration provides further reduction in ECC and improved isolation relative to the configuration of FIG. 36.

In this embodiment, antennas 3710, 3720 are planar inverted-F antennas (PIFA). The conducting segments of antennas 3710, 3720 carrying the highest current are the segments connecting the antenna to ground—segments 3711, 3712 in antenna 3710 and segments 3721, 3722 in antenna 3720, with the current in the directions indicated. Since the planar antennas are in orthogonal planes, the currents in the high-current segments of the respective antennas are likewise in orthogonal planes. This configuration effectively reduces the ECC of the two antennas.

FIG. 38 depicts an illustrative embodiment of polarization selectivity in a multi-antenna device 3800 including a multi-element antenna 3810. In this embodiment, antenna 3810 and antenna 3820 are oriented at 90° relative to each other and are in different planes; antenna 3810 is in the y-z plane while antenna 3820 is in the x-y plane. Antenna 3810 includes two antenna elements 3811, 3812 with a conducting connector 3815, in an arrangement similar to, for example, the antennas shown in FIGS. 4, 5 and 6A. In this embodiment, an antenna configuration for reduced ECC and/or increased isolation can be obtained by first determining the antenna radiation pattern for antenna 3810 (ignoring the presence of antenna 3820), and then determining the antenna pattern for antenna 3820 (ignoring the presence of antenna 3810). Antenna 3820 can then be moved and/or reoriented so that its pattern fits the nulls in the pattern of antenna 3810.

FIG. 39 depicts an illustrative embodiment of polarization selectivity in a multi-antenna device 3900 having four antennas. In this embodiment, four single-element antennas 3910, 3920, 3930, 3940 form a square array. Antennas 3910 and 3930 are at opposite corners and are in the x-y plane. Antennas 3920 and 3940 are at the other pair of opposite corners and are in the y-z and x-z planes respectively. In this embodiment, each of the four antennas is in a plane orthogonal to that of its two nearest neighbors. Since the antennas are each located at a corner of the ground plane 3601, the distance between any two antennas is the greatest available given the size of the ground plane. Accordingly, the antenna isolation is increased while the ECC is reduced.

FIG. 40 depicts an illustrative embodiment of polarization selectivity in a multi-antenna device 4000 including two multi-element antennas 4010, 4020. As shown schematically in FIG. 40, each antenna includes two antenna elements with a conducting connector. Antenna 4010 is in the y-z plane, while antenna 4020 is in the x-y plane. Since the two antennas are oriented orthogonal to each other, the antenna configuration will provide substantially the same performance in certain orientations.

FIG. 41 depicts an illustrative embodiment of polarization selectivity in a multi-antenna device 4100 having eight antennas in a square array. In this embodiment, antennas 4110 and 4150 are in the y-z plane, on opposite sides of the square (in ground plane 3601), and are oriented at 180° relative to each other. Antennas 4130 and 4170 are in the x-z plane, on opposite sides of the square, and are oriented at 180° relative to each other.(as well as orthogonal to antennas 4110, 4150). Antennas 4120, 4140, 4160, 4180 are at the corners of the square and are in the x-y plane. There are thus four pairs of antennas in which each antenna is oriented at 180° relative to its neighbor and is in a different plane. Each antenna is in a plane orthogonal to that of its two nearest neighbors. As noted above with reference to FIG. 37, this configuration is useful for reducing the ECC of the device. The isolation between any two antennas is related to the distance between them. Accordingly, each of the eight antennas has the greatest available isolation (given the size of the square) when equidistant from its two neighbors, as shown in FIG. 41. It will be appreciated that other arrangements of antennas (for example, in a circular, rectangular, or oval array) are within the scope of the disclosure.

FIG. 42 depicts an illustrative embodiment of polarization selectivity in a multi-antenna device 4200 having a symmetric arrangement of four multi-element antennas 4210, 4220, 4230, 4240. In this embodiment, antennas 4210 and 4230 are in the x-y plane while antennas 4220 and 4240 have elements in the x-z and y-z planes. In each of the multi-element antennas, one element is oriented at 180° relative to the nearest element of the neighboring antenna (e.g. elements 4231, 4241 of antennas 4230, 4240 respectively) and in a different plane. Relative to the arrangement of FIG. 41, the four antennas in FIG. 42 have greater distance between them and thus greater isolation.

FIG. 43 depicts an illustrative embodiment of polarization selectivity in a multi-antenna device 4300 having an asymmetric arrangement of four multi-element antennas 4310, 4320, 4330, 4340. In this embodiment, antennas 4310 and 4330 are in the x-y plane while antennas 4320 and 4340 are in the x-y plane. Due to the asymmetry of the antenna positions, the performance of the antenna configuration will vary with orientation of the device.

In the configurations shown in FIGS. 36-43, combinations of fixed antennas (single-element and two-element) are selected to reduce ECC and increase isolation. In another embodiment, the antennas are provided with motors so that they are movable with respect to the ground plane and each other. FIG. 44 schematically illustrates an arrangement 4400 with two antennas 4410, 4420 connected to motors 4401, 4402. The motors can move the antennas along the edge of the ground plane to alter the distance D between the antennas (where increasing D increases antenna isolation), and/or cause the antennas to rotate about the edge of the ground plane, so as to alter the angle A between the planes of the respective antennas. In an embodiment, the device can include a sensor 4451 for detecting changes in orientation of the device, and a processor 4452 that can transmit instructions to one or more of the motors to move the antennas according to the change in orientation, thereby reconfiguring the antennas to maintain antenna performance.

FIG. 45 depicts an illustrative embodiment of a method 4500 that can be applied to the subject disclosure. In step 4502, the orientation of the device is determined (e.g. using sensor 4451). A set of antennas is selected in accordance with the orientation (step 4504). The set of antennas represents an antenna configuration for the device. In an embodiment, the device has a plurality of single-element and multi-element antennas, so that a wide variety of antenna configurations can be provided.

In this embodiment, the selected set of antennas includes two antennas, either of which may be a single-element antenna or a multi-element antenna. The antenna pattern for the first antenna in the selected set is determined (step 4506). If the selected set has a single-element antenna and a multi-element antenna (as shown in FIG. 38), the antenna pattern for the multi-element antenna is determined first (step 4507). The antenna pattern for the second antenna is then determined (step 4508). If the device has a movable antenna (step 4510), that antenna is moved so that its antenna pattern fits nulls in the antenna pattern of the first antenna (step 4512).

The envelope correlation coefficient and isolation are then evaluated for the selected set of antennas (step 4514). If antenna performance is improved by using the selected set of antennas (step 4516), the selected set is designated the new antenna configuration for the device (step 4518), until a configuration is found that provides further improvement. A new set of antennas can then be selected and evaluated.

It is possible that when antennas are selected for a desirable polarity configuration in accordance with the embodiments of the subject disclosure that the ECC performance of the selected antennas can degrade as the phase of RF signals radiated by the antennas changes. To mitigate a degradation in ECC performance, a near-field electromagnetic sensor can be placed in a vicinity of each antenna of an array of antennas having different polarity configurations. The near-field electromagnetic sensor can comprise a micro-strip antenna coupled to a phase detector that can be coupled to two or more micro-strip antennas for determining a phase difference between radiated RF signals detected by the two or more micro-strip antennas. A digital output of the phase detector representative of a differential phase measurement can be presented to a controller (such as a microprocessor) for analysis. Accordingly, the controller can use two or more near-field electromagnetic sensors to measure RF signals radiated from two or more antennas of the array of antennas to determine a phase difference between.

Thus, when two or more antennas are selected for a desirable polarity between them as described by the subject disclosure, the controller can use the near-field electromagnetic sensors in a vicinity of the two or more antennas selected. If the controller determines that the phase between the RF signals radiated by the antennas selected is such that their respective phases are additive, then a degradation in the ECC performance of the antennas can be detected by the controller. When such a condition is detected, the controller can adjust the phase of one or more of the affected antennas by directing one or more phase shifters to adjust the phase of one or more corresponding transmit signals directed to the affected antennas. Upon adjusting the phase of the one or more of the transmit signals, the controller can once again measure from the two or more near-field electromagnetic sensors the phase of the radiated signals of the selected antennas. If their respective phases have become, at least in part, subtractive (or at least partially nullifying), then the controller can make a determination that the ECC performance of the antennas has improved. Such a closed loop analysis can continue until the controller determines the phase differential of the radiated RF signals of the selected antennas is sufficiently subtractive to achieve a desirable ECC performance. In an ideal sense, the controller can continue phase adjustments until the RF radiated signals between the antennas are 180 degrees out of phase with each other. Such an ideal phase difference, however, may not always be achievable. Accordingly, the controller can be configured to cease adjustments when a differential phase threshold (e.g., 60% of an ideal phase difference between the radiated RF signals) is achieved.

FIG. 46 depicts an illustrative embodiment of a communication device 4600. The communication device 4600 can comprise a wireline and/or wireless transceiver 4602 (herein transceiver 4602), a user interface (UI) 4604, a power supply 4614, a location receiver 4616, a motion sensor 4618, an orientation sensor 4620, and a controller 4606 for managing operations thereof. The transceiver 4602 can support short-range or long-range wireless access technologies such as Bluetooth, ZigBee, WiFi, DECT, or cellular communication technologies, just to mention a few. Cellular technologies can include, for example, CDMA-1X, UMTS/HSDPA, GSM/GPRS, TDMA/EDGE, EV/DO, WiMAX, SDR, LTE, as well as other next generation wireless communication technologies as they arise. The transceiver 4602 can also be adapted to support circuit-switched wireline access technologies (such as PSTN), packet-switched wireline access technologies (such as TCP/IP, VoIP, etc.), and combinations thereof. The transceiver 4602 can be adapted to utilize any of the aforementioned adaptive antenna embodiments described above singly or in any combination.

The UI 4604 can include a depressible or touch-sensitive keypad 4608 with a navigation mechanism such as a roller ball, a joystick, a mouse, or a navigation disk for manipulating operations of the communication device 4600. The keypad 4608 can be an integral part of a housing assembly of the communication device 4600 or an independent device operably coupled thereto by a tethered wireline interface (such as a USB cable) or a wireless interface supporting for example Bluetooth. The keypad 4608 can represent a numeric keypad commonly used by phones, and/or a QWERTY keypad with alphanumeric keys. The UI 4604 can further include a display 4610 such as monochrome or color LCD (Liquid Crystal Display), OLED (Organic Light Emitting Diode) or other suitable display technology for conveying images to an end user of the communication device 4600. In an embodiment where the display 4610 is touch-sensitive, a portion or all of the keypad 4608 can be presented by way of the display 4610 with navigation features.

The display 4610 can use touch screen technology to also serve as a user interface for detecting user input. As a touch screen display, the communication device 4600 can be adapted to present a user interface with graphical user interface (GUI) elements that can be selected by a user with a touch of a finger. The touch screen display 4610 can be equipped with capacitive, resistive or other forms of sensing technology to detect how much surface area of a user's finger has been placed on a portion of the touch screen display. This sensing information can be used to control the manipulation of the GUI elements or other functions of the user interface. The display 4610 can be an integral part of the housing assembly of the communication device 4600 or an independent device communicatively coupled thereto by a tethered wireline interface (such as a cable) or a wireless interface.

The UI 4604 can also include an audio system 4612 that utilizes audio technology for conveying low volume audio (such as audio heard in proximity of a human ear) and high volume audio (such as speakerphone for hands free operation). The audio system 4612 can further include a microphone for receiving audible signals of an end user. The audio system 4612 can also be used for voice recognition applications. The UI 4604 can further include an image sensor 4613 such as a charged coupled device (CCD) camera for capturing still or moving images.

The power supply 4614 can utilize common power management technologies such as replaceable and rechargeable batteries, supply regulation technologies, and/or charging system technologies for supplying energy to the components of the communication device 4600 to facilitate long-range or short-range portable applications. Alternatively, or in combination, the charging system can utilize external power sources such as DC power supplied over a physical interface such as a USB port or other suitable tethering technologies.

The location receiver 4616 can utilize location technology such as a global positioning system (GPS) receiver capable of assisted GPS for identifying a location of the communication device 4600 based on signals generated by a constellation of GPS satellites, which can be used for facilitating location services such as navigation. The motion sensor 4618 can utilize motion sensing technology such as an accelerometer, a gyroscope, or other suitable motion sensing technology to detect motion of the communication device 4600 in three-dimensional space. The orientation sensor 4620 can utilize orientation sensing technology such as a magnetometer to detect the orientation of the communication device 4600 (north, south, west, and east, as well as combined orientations in degrees, minutes, or other suitable orientation metrics).

The communication device 4600 can use the transceiver 4602 to also determine a proximity to a cellular, WiFi, Bluetooth, or other wireless access points by sensing techniques such as utilizing a received signal strength indicator (RSSI) and/or signal time of arrival (TOA) or time of flight (TOF) measurements. The controller 4606 can utilize computing technologies such as a microprocessor, a digital signal processor (DSP), programmable gate arrays, application specific integrated circuits, and/or a video processor with associated storage memory such as Flash, ROM, RAM, SRAM, DRAM or other storage technologies for executing computer instructions, controlling, and processing data supplied by the aforementioned components of the communication device 400.

Other components not shown in FIG. 46 can be used in one or more embodiments of the subject disclosure. For instance, the communication device 4600 can include a reset button (not shown). The reset button can be used to reset the controller 4606 of the communication device 4600. In yet another embodiment, the communication device 4600 can also include a factory default setting button positioned, for example, below a small hole in a housing assembly of the communication device 4600 to force the communication device 4600 to re-establish factory settings. In this embodiment, a user can use a protruding object such as a pen or paper clip tip to reach into the hole and depress the default setting button. The communication device 400 can also include a slot for adding or removing an identity module such as a Subscriber Identity Module (SIM) card. SIM cards can be used for identifying subscriber services, executing programs, storing subscriber data, and so forth.

The communication device 4600 as described herein can operate with more or less of the circuit components shown in FIG. 46. These variant embodiments can be used in one or more embodiments of the subject disclosure.

It should be understood that devices described in the exemplary embodiments can be in communication with each other via various wireless and/or wired methodologies. The methodologies can be links that are described as coupled, connected and so forth, which can include unidirectional and/or bidirectional communication over wireless paths and/or wired paths that utilize one or more of various protocols or methodologies, where the coupling and/or connection can be direct (e.g., no intervening processing device) and/or indirect (e.g., an intermediary processing device such as a router).

FIG. 47 depicts an exemplary diagrammatic representation of a machine in the form of a computer system 4700 within which a set of instructions, when executed, may cause the machine to perform any one or more of the embodiments described above. One or more instances of the machine can utilize the aforementioned adaptive antenna embodiments singly or in any combination. In some embodiments, the machine may be connected (e.g., using a network 4726) to other machines. In a networked deployment, the machine may operate in the capacity of a server or a client user machine in server-client user network environment, or as a peer machine in a peer-to-peer (or distributed) network environment.

The machine may comprise a server computer, a client user computer, a personal computer (PC), a tablet PC, a smart phone, a laptop computer, a desktop computer, a control system, a network router, switch or bridge, or any machine capable of executing a set of instructions (sequential or otherwise) that specify actions to be taken by that machine. It will be understood that a communication device of the subject disclosure includes broadly any electronic device that provides voice, video or data communication. Further, while a single machine is illustrated, the term “machine” shall also be taken to include any collection of machines that individually or jointly execute a set (or multiple sets) of instructions to perform any one or more of the methods discussed herein.

The computer system 4700 may include a processor (or controller) 4702 (e.g., a central processing unit (CPU), a graphics processing unit (GPU, or both), a main memory 4704 and a static memory 4706, which communicate with each other via a bus 4708. The computer system 4700 may further include a display unit 4710 (e.g., a liquid crystal display (LCD), a flat panel, or a solid state display. The computer system 4700 may include an input device 4712 (e.g., a keyboard), a cursor control device 4714 (e.g., a mouse), a disk drive unit 4716, a signal generation device 4718 (e.g., a speaker or remote control) and a network interface device 4720. In distributed environments, the embodiments described in the subject disclosure can be adapted to utilize multiple display units 4710 controlled by two or more computer systems 4700. In this configuration, presentations described by the subject disclosure may in part be shown in a first of the display units 4710, while the remaining portion is presented in a second of the display units 4710.

The disk drive unit 4716 may include a tangible computer-readable storage medium 4722 on which is stored one or more sets of instructions (e.g., software 4724) embodying any one or more of the methods or functions described herein, including those methods illustrated above. The instructions 4724 may also reside, completely or at least partially, within the main memory 4704, the static memory 4706, and/or within the processor 4702 during execution thereof by the computer system 4700. The main memory 4704 and the processor 4702 also may constitute tangible computer-readable storage media.

Dedicated hardware implementations including, but not limited to, application specific integrated circuits, programmable logic arrays and other hardware devices that can likewise be constructed to implement the methods described herein. Application specific integrated circuits and programmable logic array can use downloadable instructions for executing state machines and/or circuit configurations to implement embodiments of the subject disclosure. Applications that may include the apparatus and systems of various embodiments broadly include a variety of electronic and computer systems. Some embodiments implement functions in two or more specific interconnected hardware modules or devices with related control and data signals communicated between and through the modules, or as portions of an application-specific integrated circuit. Thus, the example system is applicable to software, firmware, and hardware implementations.

In accordance with various embodiments of the subject disclosure, the operations or methods described herein are intended for operation as software programs or instructions running on or executed by a computer processor or other computing device, and which may include other forms of instructions manifested as a state machine implemented with logic components in an application specific integrated circuit or field programmable gate array. Furthermore, software implementations (e.g., software programs, instructions, etc.) including, but not limited to, distributed processing or component/object distributed processing, parallel processing, or virtual machine processing can also be constructed to implement the methods described herein. It is further noted that a computing device such as a processor, a controller, a state machine or other suitable device for executing instructions to perform operations or methods may perform such operations directly or indirectly by way of one or more intermediate devices directed by the computing device.

While the tangible computer-readable storage medium 4722 is shown in an example embodiment to be a single medium, the term “tangible computer-readable storage medium” should be taken to include a single medium or multiple media (e.g., a centralized or distributed database, and/or associated caches and servers) that store the one or more sets of instructions. The term “tangible computer-readable storage medium” shall also be taken to include any non-transitory medium that is capable of storing or encoding a set of instructions for execution by the machine and that cause the machine to perform any one or more of the methods of the subject disclosure.

The term “tangible computer-readable storage medium” shall accordingly be taken to include, but not be limited to: solid-state memories such as a memory card or other package that houses one or more read-only (non-volatile) memories, random access memories, or other re-writable (volatile) memories, a magneto-optical or optical medium such as a disk or tape, or other tangible media which can be used to store information. Accordingly, the disclosure is considered to include any one or more of a tangible computer-readable storage medium, as listed herein and including art-recognized equivalents and successor media, in which the software implementations herein are stored.

Although the present specification describes components and functions implemented in the embodiments with reference to particular standards and protocols, the disclosure is not limited to such standards and protocols. Each of the standards for Internet and other packet switched network transmission (e.g., TCP/IP, UDP/IP, HTML, HTTP) represent examples of the state of the art. Such standards are from time-to-time superseded by faster or more efficient equivalents having essentially the same functions. Wireless standards for device detection (e.g., RFID), short-range communications (e.g., Bluetooth, WiFi, Zigbee), and long-range communications (e.g., WiMAX, GSM, CDMA, LTE) can be used by computer system 4700.

The illustrations of embodiments described herein are intended to provide a general understanding of the structure of various embodiments, and they are not intended to serve as a complete description of all the elements and features of apparatus and systems that might make use of the structures described herein. Many other embodiments will be apparent to those of skill in the art upon reviewing the above description. The exemplary embodiments can include combinations of features and/or steps from multiple embodiments. Other embodiments may be utilized and derived therefrom, such that structural and logical substitutions and changes may be made without departing from the scope of this disclosure. Figures are also merely representational and may not be drawn to scale. Certain proportions thereof may be exaggerated, while others may be minimized. Accordingly, the specification and drawings are to be regarded in an illustrative rather than a restrictive sense.

Although specific embodiments have been illustrated and described herein, it should be appreciated that any arrangement calculated to achieve the same purpose may be substituted for the specific embodiments shown. This disclosure is intended to cover any and all adaptations or variations of various embodiments. Combinations of the above embodiments, and other embodiments not specifically described herein, can be used in the subject disclosure.

The Abstract of the Disclosure is provided with the understanding that it will not be used to interpret or limit the scope or meaning of the claims. In addition, in the foregoing Detailed Description, it can be seen that various features are grouped together in a single embodiment for the purpose of streamlining the disclosure. This method of disclosure is not to be interpreted as reflecting an intention that the claimed embodiments require more features than are expressly recited in each claim. Rather, as the following claims reflect, inventive subject matter lies in less than all features of a single disclosed embodiment. Thus the following claims are hereby incorporated into the Detailed Description, with each claim standing on its own as a separately claimed subject matter.

It is to be understood that although the disclosure has been described above in terms of particular embodiments, the foregoing embodiments are provided as illustrative only, and do not limit or define the scope of the disclosure.

Various other embodiments, including but not limited to the following, are also within the scope of the claims. For example, the elements or components of the various multimode antenna structures described herein may be further divided into additional components or joined together to form fewer components for performing the same functions. For example, the antenna elements and the connecting element or elements that are part of a multimode antenna structure may be combined to form a single radiating structure having multiple feed points operatively coupled to a plurality of antenna ports or feed points.

It is further noted that the low band and high band antennae structures described in the subject disclosure may be different or dissimilar antenna types, such as, for example, monopole, PIFA, loop, dielectric or other structures known in the art. It is also noted that the embodiments described herein may represent other sub-frequency ranges such as, for example, low band, mid band, and high band. Accordingly, the antenna structures described herein may have differing antenna types, and differing frequency ranges.

Having described embodiments of the present disclosure, it should be apparent that modifications can be made without departing from the spirit and scope of the disclosure. 

What is claimed is:
 1. A method comprising: determining, by a communication device comprising a processor, a usage state of the communication device, wherein the communication device comprises a plurality of selectable antennas for engaging in wireless communications, the usage state including an orientation of the communication device; selecting, by the communication device, a set of antennas from the plurality of antennas in accordance with the usage state; obtaining, by the communication device, an antenna gain pattern for an antenna in the selected set of antennas; evaluating, by the communication device, an expected performance of an antenna configuration corresponding to the selected set of antennas relative to a performance of an existing antenna configuration; and initiating, by the communication device, usage of the antenna configuration corresponding to the selected set of antennas in accordance with improved performance relative to the existing antenna configuration, wherein the antenna configuration comprises a polarization configuration, and wherein the selected set of antennas comprises a first antenna element in a first plane and a second antenna element in a different second plane.
 2. The method of claim 1, wherein the selected set of antennas comprises a first antenna and a second antenna, the first antenna including the first antenna element, the second antenna element, and a connector connecting the first antenna element and the second antenna element, wherein the obtaining comprises obtaining a first antenna gain pattern for the first antenna and obtaining a second antenna gain pattern for the second antenna, and wherein the second antenna gain pattern fits a null in the first antenna gain pattern.
 3. The method of claim 1, wherein the evaluating further comprises evaluating a first envelope correlation coefficient for the existing antenna configuration and a second envelope correlation coefficient for the antenna configuration corresponding to the selected set of antennas.
 4. The method of claim 1, wherein the evaluating further comprises evaluating a first antenna isolation for the existing antenna configuration and a second antenna isolation for the antenna configuration corresponding to the selected set of antennas.
 5. The method of claim 1, wherein the first antenna element and the second antenna element are oriented at 180° relative to each other, and wherein the first plane and the second plane are orthogonal.
 6. The method of claim 5, wherein the selected set of antennas further comprises a third antenna element and a fourth antenna element, the first, second, third and fourth antenna elements being arranged in an array with the first antenna element and the third antenna element diagonally opposite and substantially coplanar, and wherein the fourth antenna is in a plane orthogonal to the first plane and orthogonal to the second plane.
 7. The method of claim 1, wherein the first antenna element comprises a first antenna segment connected to a ground plane and having a first current therein, wherein the second antenna element comprises a second antenna segment connected to the ground plane and having a second current therein, wherein the first current and the second current are in planes orthogonal to each other, and wherein the antenna configuration has a reduced envelope correlation coefficient relative to the existing antenna configuration.
 8. The method of claim 1, wherein a first antenna of the selected set of antennas is provided with a motor for moving the first antenna relative to a second antenna of the selected set of antennas, and further comprising directing the motor to move the first antenna relative to the second antenna so that the first antenna and the second antenna are in different planes.
 9. The method of claim 1, further comprising: measuring electric fields associated with the selected set of antennas while in use and their corresponding phase difference; determining whether the fields are additive based on the phase difference; and responsive to a determination that the fields are additive, adjusting electric fields of at least one antenna of the selected set of antennas to cause the fields to be subtractive.
 10. The method of claim 1, wherein the determining further comprises detecting a change in the orientation of the communication device between a vertical orientation and a horizontal orientation.
 11. An antenna structure for a communication device, comprising: a set of antennas selected from a plurality of antennas of the communication device, the set of antennas being selected in accordance with a usage state of the communication device; and a controller that facilitates selection of the set of antennas based on an evaluation of a performance of an antenna configuration corresponding to the selected set of antennas indicating an improved performance relative to a previous antenna configuration, wherein the evaluation comprises an envelope correlation coefficient for the antenna configuration corresponding to the selected set of antennas and an antenna isolation for the antenna configuration corresponding to the selected set of antennas.
 12. The antenna structure of claim 11, wherein the selected set of antennas comprises a first antenna and a second antenna, the first antenna including a first antenna element, a second antenna element, and a connector connecting the first antenna element and the second antenna element, wherein the evaluation is performed based on a first antenna gain pattern for the first antenna and a second antenna gain pattern for the second antenna, and wherein the second antenna gain pattern fits a null in the first antenna gain pattern.
 13. The antenna structure of claim 11, wherein the selected set of antennas comprises a first antenna and a second antenna, wherein the first antenna is in a first plane and the second antenna is in a different second plane, wherein a first antenna element of the first antenna and a second antenna element of the second antenna are oriented at 180° relative to each other, and wherein the first plane and the second plane are orthogonal.
 14. The antenna structure of claim 11, wherein the selected set of antennas comprises a first antenna and a second antenna, and further comprising a motor coupled to the first antenna for moving the first antenna relative to the second antenna.
 15. The antenna structure of claim 14, further comprising: a sensor for detecting an orientation of the communication device; and a processor coupled to the sensor and to the motor for directing the motor to move the first antenna in accordance with a change in the orientation, wherein the processor directs the motor to move the first antenna relative to the second antenna so that the first antenna and the second antenna are in different planes.
 16. The antenna structure of claim 11, further comprising: a plurality of near-field sensors, each located in a vicinity of the plurality of antennas for measuring a phase differential between the set of antennas selected; and a phase shifter for adjusting radiated signals from at least one of the set of antennas when the phase differential between the set of antennas is additive.
 17. A machine-readable storage medium comprising instructions, wherein responsive to executing the instructions, a processor performs operations comprising: detecting a usage state of a communication device, wherein the communication device comprises a plurality of selectable antennas for engaging in wireless communications, the usage state including an orientation of the communication device; selecting a set of antennas from the plurality of antennas in accordance with the usage state; obtaining a first antenna gain pattern for a first antenna in the selected set of antennas; obtaining a second antenna gain pattern for a second antenna in the selected set of antennas; and evaluating an expected performance of an antenna configuration corresponding to the selected set of antennas relative to a performance of an existing antenna configuration, wherein the evaluating comprises determining an envelope correlation coefficient for the antenna configuration corresponding to the selected set of antennas and an antenna isolation for the antenna configuration corresponding to the selected set of antennas, wherein the evaluating is based on the first antenna gain pattern and the second antenna gain pattern, and wherein the second antenna gain pattern fits a null in the first antenna gain pattern.
 18. The machine-readable storage medium of claim 17, wherein the first antenna is in a first plane and the second antenna is in a different second plane, wherein the first antenna and the second antenna respectively include a first antenna element and a second antenna element, and wherein the first antenna element and the second antenna element are oriented at 180° relative to each other.
 19. The machine-readable storage medium of claim 17, wherein the first antenna is provided with a motor for moving the first antenna relative to the second antenna.
 20. The machine-readable storage medium of claim 19, wherein the operations further comprise directing the motor to move the first antenna relative to the second antenna so that the first antenna and the second antenna are in different planes. 